Test mode and test method for a temperature tamper detection circuit

ABSTRACT

An integrated circuit temperature sensor includes a sensing circuit operable to determine whether the integrated circuit is currently exposed to one of a relatively low temperature or a relatively high temperature. A selection circuit operates to select a measured voltage across the base-emitter of a bipolar transistor of the integrated circuit if the sensing circuit indicates that the integrated circuit is currently exposed to the relatively low temperature or, alternatively, select a measured delta voltage across the base-emitter of the bipolar transistor of the integrated circuit if the sensing circuit indicates that the integrated circuit is currently exposed to the relatively high temperature. A comparator then compares the selected measured voltage across the base-emitter of the bipolar transistor against a first reference voltage indicative of a too cold temperature condition or compares the selected measured delta voltage across the base-emitter of the bipolar transistor against a second reference voltage indicative of a too hot temperature condition. As a result of the comparison, detection may be made as to whether the integrated circuit is currently exposed to one of either a too cold or too hot temperature. In a test mode, the circuit is exposed to a readily available temperature, such as room temperature, and the measured delta voltage across the base-emitter and/or the measured voltage across the base-emitter are scaled in accordance with that available temperature for application to the comparator. Alternatively, in test mode the reference voltages are scaled to intersect with the measured delta voltage across the base-emitter and/or the measured voltage across the base-emitter at the available temperature.

PRIORITY CLAIM

This application claims the benefit of U.S. Provisional Application for Patent Ser. No. 60/739,150, filed Nov. 22, 2005, the disclosure of which is hereby incorporated by reference.

BACKGROUND OF THE INVENTION

1. Technical Field of the Invention

The present invention relates to temperature alarm circuitry, and more particularly to a test mode and method useful with respect to a temperature alarm circuit and a tamper detection circuit.

2. Description of Related Art

There exist electronics systems in which very sensitive data may be stored in an integrated circuit (IC). For example, some portable credit card scanners store credit card data in volatile memory which, if pilfered, would potentially expose credit card holders to credit card fraud.

In an effort to prevent such pilfering of data, techniques exist for detecting when an IC or system is being tampered with, and destroying the stored data in response to the detection. In this way, sensitive data stored in such systems remains inaccessible.

One known way of attempting to improperly access stored data is to subject the IC or system to changes in temperature, for example by exposing the IC or system to extreme heat or extreme cold. There is accordingly a need for a type of tamper detection which involves detecting whether the temperature of the IC or system falls outside an expected operating range (for example, too hot or too cold in comparison to certain temperature thresholds, wherein the expected operating range falls between the high and low thresholds), and asserting an alarm responsive to such detection. Such a temperature tamper detector would serve the purpose of preventing a tamperer from accessing stored, sensitive data in part by the tamperer changing the operating temperature of the IC or system.

The best way to test operation of the temperature tamper detector is to subject the detector itself, or the IC or system within which the detector is installed, to a temperature which is outside the expected operating range (i.e., an extreme temperature). If the detector output is monitored and it changes state in response to the applied extreme temperature, then one can assume that the temperature tamper detector has been programmed and is operating properly. This testing solution, however, is not particularly practical or economical since it necessitates the generation of an extreme temperature (too hot or too cold) outside the expected operating range followed by the exposure of the detector, IC or system to that extreme temperature. In some testing instances it may be impossible or inconvenient to generate the required extreme temperature for performing the test. There is a need in the art to be able to test operation of the temperature tamper detector at a more readily available and convenient temperature, such as room temperature, which lies within the expected operating temperature range.

SUMMARY OF THE INVENTION

In accordance with an embodiment of the invention, an integrated circuit temperature sensor comprises a scaling circuit operable to scale a voltage across the base-emitter of a bipolar transistor of the integrated circuit and a delta voltage across the base-emitter of a bipolar transistor of the integrated circuit to lie within a voltage reference range less than a supply voltage when the sensor is exposed to a temperature within an expected operating temperature range. A first multiplexer having first and second inputs is coupled to receive the scaled voltage across the base-emitter of the bipolar transistor of the integrated circuit and the scaled delta voltage across the base-emitter of the bipolar transistor of the integrated circuit, the first multiplexer selecting one of the scaled voltages at the first and second inputs for output responsive to a test mode signal. A second comparator operates to compare the voltage output from the first multiplexer against a reference voltage and generate an output signal responsive thereto.

An integrated circuit temperature sensing method comprises scaling a voltage across the base-emitter of a bipolar transistor of the integrated circuit and a delta voltage across the base-emitter of a bipolar transistor of the integrated circuit when in test mode to lie within a voltage reference range less than a supply voltage while the sensor is exposed to a temperature within an expected operating temperature range. The scaled voltage across the base-emitter of the bipolar transistor of the integrated circuit is selected if programming a high temperature threshold. Alternatively, the scaled delta voltage across the base-emitter of the bipolar transistor of the integrated circuit is selected if programming a low temperature threshold. The selected voltage is then compared against a reference voltage to generate an output signal.

In accordance with another embodiment, an integrated circuit temperature sensor comprises a reference voltage generator operable in a test mode to generate an incrementing reference voltage within a certain voltage range. A circuit operates during test mode to scale a temperature dependent voltage signal concerning the base-emitter of a bipolar transistor of the integrated circuit and which is indicative of circuit operation outside of an acceptable temperature range to lie within the certain voltage range when the integrated circuit is exposed to a temperature that is within the acceptable temperature range. A comparator then compares the scaled temperature dependent voltage signal against the incrementing reference voltage to generate an output signal and identify a certain incremented reference voltage at which the output signal changes state.

In accordance with another embodiment, an integrated circuit temperature sensor comprises a reference voltage generator operable in a test mode to generate an incrementing reference voltage signal within a certain voltage range. A scaling circuit scales a temperature dependent voltage signal concerning the base-emitter of a bipolar transistor of the integrated circuit to lie within the certain voltage range less than a supply voltage when the sensor is exposed to a temperature within an expected operating temperature range. A comparator then compares the incrementing reference voltage signal against the scaled temperature dependent voltage signal and generate an output signal responsive thereto.

In accordance with an embodiment, an integrated circuit temperature sensor comprises a scaling circuit operable to selectively shift a voltage reference from a first range less than a supply voltage for non-test mode operation to a second range less than a supply voltage during test mode operation such that a voltage across the base-emitter of a bipolar transistor of the integrated circuit and a delta voltage across the base-emitter of a bipolar transistor of the integrated circuit lie within the second voltage reference range during the test mode of operation when the sensor is exposed to a temperature within an expected operating temperature range. A multiplexer includes first and second inputs coupled to receive the voltage across the base-emitter of the bipolar transistor of the integrated circuit and the delta voltage across the base-emitter of the bipolar transistor of the integrated circuit. The multiplexer selects one of the scaled voltages at the first and second inputs for output. A comparator is operable to compare the voltage output from the multiplexer against the shifted reference voltage and generate an output signal responsive thereto.

In accordance with another embodiment, an integrated circuit temperature sensing method comprises selectively shifting a voltage reference from a first range less than a supply voltage for non-test mode operation to a second range less than a supply voltage during test mode operation such that a voltage across the base-emitter of a bipolar transistor of the integrated circuit and a delta voltage across the base-emitter of a bipolar transistor of the integrated circuit to lie within the second voltage reference range during the test mode of operation when the sensor is exposed to a temperature within an expected operating temperature range; selecting one of the voltage across the base-emitter of the bipolar transistor of the integrated circuit and the delta voltage across the base-emitter of the bipolar transistor of the integrated circuit for output; and comparing the selected voltage against the shifted reference voltage to generate an output signal.

In accordance with another embodiment, an integrated circuit temperature sensor comprises a reference voltage generator operable in a test mode to generate an incrementing reference voltage within a first voltage range less than a supply voltage. A circuit is operable to selectively shift the incrementing voltage reference from the first voltage range to a second range less than a supply voltage during test mode operation such that a temperature dependent voltage signal concerning the base-emitter of a bipolar transistor of the integrated circuit and which is indicative of circuit operation outside of an acceptable temperature range lies within the second voltage range when the integrated circuit is exposed to a temperature that is within the acceptable temperature range. A comparator is operable to compare the temperature dependent voltage signal against the shifted incrementing reference voltage to generate an output signal and identify a certain incremented reference voltage at which the output signal changes state.

BRIEF DESCRIPTION OF THE DRAWINGS

A more complete understanding of the method and apparatus of the present invention may be acquired by reference to the following Detailed Description when taken in conjunction with the accompanying Drawings wherein:

FIGS. 1A and 1B are graphs illustrating Vbe and delta Vbe over an exemplary temperature range, and show the scaling of Vbe and delta Vbe and the scaling of Vref, respectively;

FIGS. 2A and 2B are block diagrams of the temperature tampering detection circuit according to embodiments of the present invention;

FIG. 3A shows a circuit diagram of a possible implementation of a comparator for the temperature tampering detection circuit;

FIG. 3B shows a circuit diagram of a possible implementation of a digital-to-analog converter circuit for the temperature tampering detection circuit;

FIG. 3C shows a circuit diagram of a possible implementation of a counter for the temperature tampering detection circuit which is capable of incrementing and decrementing;

FIG. 3D shows a circuit diagram of a possible implementation of the signal generator circuits for the temperature tampering detection circuit;

FIG. 3E shows a circuit diagram of a possible implementation of a multiplexer circuit for the temperature tampering detection circuit;

FIG. 3F shows a circuit diagram of a possible implementation of a hysteresis circuit for the temperature tampering detection circuit;

FIG. 3G shows a diagram for a possible implementation of a decoder circuit for the temperature tampering detection circuit;

FIG. 3H shows a diagram for a possible implementation of a 4:16 decoder for the decoder circuit of FIG. 3G;

FIG. 3I shows a diagram for a possible implementation of a 2:4 decoder for the decoder circuit of FIG. 3G;

FIG. 3J shows a circuit diagram for a possible implementation of resistor array for the temperature tampering detection circuit;

FIG. 3K shows a circuit diagram for a possible implementation of a bandgap voltage generator for the temperature tampering detection circuit; and

FIG. 3L shows a circuit diagram for a possible implementation of a signal generator circuit for the temperature tampering detection circuit.

DETAILED DESCRIPTION OF THE DRAWINGS

One type of tamper detection involves detecting whether the temperature of the IC or system falls outside (too high or too low) an expected operating range, and asserting an alarm responsive to such detection. Temperature detection serves to prevent a tamperer from accessing stored, sensitive data in part by the tamperer changing the operating temperature of the IC or system.

It is recognized that a substantially linear relationship exists between temperature and the change in the base-emitter voltage of a bipolar transistor (hereinafter “delta Vbe”) within the IC or system to be protected. Specifically, delta Vbe is substantially process insensitive and can be used for temperature sensing due to its superior linearity. To achieve good voltage/temperature sensitivity, the delta Vbe voltage needs to be amplified from its typical voltage variation over normal temperature conditions.

One difficulty encountered with monitoring delta Vbe over the entire operating temperature range is that the variation of amplified delta Vbe may not always be within the operating temperature range of the Vcc voltage supply to the IC or system. For example, with delta Vbe sensitivity of 6 mv/degree C, the Vbe voltage may vary from −1.2 v at −45 degrees C. to 2.4 v at 120 degrees C. Taking into consideration processing variations, this 1.2 v delta Vbe variation between low and high temperature fails to fit within some relatively small Vcc operating ranges for the IC or system which may have a minimum Vcc voltage level of 1.2 v.

In an effort to ensure that temperature is accurately sensed over the entire operating temperature range of the IC, an embodiment of the present invention not only monitors delta Vbe but also Vbe. Specifically, Vbe, the base-emitter voltage of a bipolar transistor within the IC or system, also varies substantially linearly with temperature. Whereas delta Vbe varies in direct proportion to changes in temperature, Vbe varies in indirect proportion to temperature. A graph comparing Vbe and delta Vbe over an exemplary operational temperature range for the IC or system is shown in FIG. 1.

In order to monitor temperature over an entire operating temperature range, such as −40 degrees C. to 125 degrees C., an embodiment of the present invention monitors delta Vbe at relatively low temperatures and Vbe at elevated, or relatively high, temperatures. Specifically, delta Vbe is monitored at relatively low temperatures to determine whether the temperature falls below a first temperature (first threshold), and Vbe is monitored at relatively high temperatures to determine whether the temperature rises above a second temperature (second threshold). Because delta Vbe is less than a Vcc supply voltage level at relatively lower temperatures and Vbe is less than the Vcc supply voltage level at relatively higher temperatures, delta Vbe and Vbe may be monitored in this way by relatively easily comparing each to one or more reference voltages, such as a selected stable reference voltage Vref (see, FIG. 1) that, for example, is based upon a bandgap voltage. By comparing delta Vbe with the selected reference voltage Vref when the IC or system is exposed to a lower or relatively low temperature and by comparing Vbe with the selected reference voltage Vref when the IC or system is exposed to a higher or relatively high temperature, the temperature tamper detect circuit of the present invention is capable of detecting tampering activity occurring as a result of altering the temperature outside of an expected operating temperature range. Advantageously, the circuit of the present invention is suitable for operation to make too low and/or too high temperature detections even when the IC or system is being operated at relatively low Vcc supply voltage levels (for example, 1.2 v or 1.8 v, and see FIG. 1 at lower than 2.4 v).

FIGS. 2A and 2B are block diagrams of the temperature tampering detection circuit (TTDC) 10 according to embodiments of the present invention. TTDC 10 is provided within an IC or system 11 and includes a first comparator 12 which compares delta Vbe with Vbe and generates an output signal that is used to determine whether delta Vbe or Vbe is to be compared to a reference voltage Vref. The comparison of delta Vbe with Vbe enables the circuit 10 to determine whether the IC or system 11 is currently experiencing a relatively low or relatively high temperature environment (see, FIG. 1 which illustrates the relationship between delta Vbe and Vbe over a range of temperatures, where temperatures less than the temperature at the intersection point are relatively low, while temperatures more than the temperature at the intersection point are relatively high). As will be discussed later in more detail, the comparison of one of delta Vbe or Vbe to the reference voltage Vref is being made in order to determine whether a temperature “out of range” condition exists for the IC or system (i.e., whether the IC or system is experiencing either a too cold or too hot environment in comparison to corresponding temperature thresholds). Comparator 12 may include hysteresis in order to avoid the output of comparator 12 from undesirably oscillating during the comparison operation.

TTDC 10 further includes multiplexer circuitry 14 having data inputs coupled to receive delta Vbe and Vbe signals and a select input coupled to the output of comparator 12. Based on the select input state (corresponding to relatively low/cold or relatively high/hot), one of the delta Vbe and Vbe signals is selected by multiplexer circuitry 14 for output from the multiplexer circuitry 14. More specifically, if the select input state corresponds to relatively low/cold, then multiplexer selects the delta Vbe signal at the data input since FIG. 1 shows that this signal is better used for making the too cold determination. Alternatively, if the select input state corresponds to relatively high/hot, then multiplexer selects the Vbe signal at the data input since FIG. 1 shows that this signal is better used for making the too hot determination.

TTDC 10 further includes a comparator 16 having its data inputs coupled to receive both a first reference voltage Vref and the selected output of multiplexer circuitry 14 (either delta Vbe or Vbe). Comparator 12, which determines whether delta Vbe or Vbe is to be utilized in determining whether an “out of range” temperature exists, thus is used to provide the appropriate delta Vbe or Vbe signal to comparator 16 through controlling the selection operation of the multiplexer circuit 14. The output of comparator 16 changes state based on the result of the comparison operation. Thus, if in relatively low mode, comparator 16 changes state when Vbe is less than a selected vref as this would be indicative of a too cold condition (see, reference 23 in FIG. 1). Conversely, if in relatively high mode, comparator 16 changes state when delta Vbe is less than a selected vref as this would be indicative of a too hot condition (see, reference 21 in FIG. 1).

The output from the comparator 16 may be an alarm signal temp_alrm (possibly buffered in a flip-flop FF). The alarm signal, when asserted, may cause the IC or system 11 to take measures to prevent sensitive data from being copied. For example, the alarm signal may cause the IC or system 11 to act on the stored, sensitive data by taking actions to erase and/or corrupt the stored data. The comparator 16 may have hysteresis in order to avoid the output of comparator 16 from undesirably oscillating during the comparison operation.

In order to monitor delta Vbe at low temperatures and Vbe at elevated temperatures, in one embodiment of the invention two voltage references Vref are utilized: one voltage reference Vref1 for use in determining whether the temperature falls below a predetermined low temperature (for the comparator 16 operation in determining whether delta Vbe is less than the voltage threshold), and a second voltage reference Vref2 for use in determining whether the temperature rises above a predetermined high temperature (for the comparator 16 operation in determining whether Vbe is less than the voltage threshold). With reference to FIGS. 2A and 2B, TTDC 10 further includes a first signal generator circuit 18 which generates a digital signal corresponding to first voltage reference Vref1, and a second signal generator circuit 20 which generates a digital signal corresponding to second voltage reference Vref2.

It will be understood, with reference to FIG. 1, that only a single vref value is needed as that specified vref could be used by the comparator 16 for purposes of making either of the temperature too low or temperature too high determinations. In such a case, only a single signal generator circuit 18/20 would be needed.

However, in a situation where a single voltage reference vref would not intersect the delta Vbe and Vbe lines at appropriate low and high temperature locations, respectively, of interest, it would be preferred to use two voltage references Vref selectively chosen for the comparison operation based on whether the IC or system 11 was relatively cold or relatively hot.

First signal generator circuit 18 and second signal generator circuit 20 may each include programmable circuitry for setting and/or programming the digital signals corresponding to reference voltages Vref. Such programmability allows for the digital signals representing the reference voltages to be programmed to customer requirements. For example, first and second signal generator circuits 18, 20 may include fuses representing digital values which are selectively blown (for example, by laser) to set a digital value in circuits 18, 20 which corresponds to a certain voltage reference for use in the comparison operation performed by comparator 16. Alternatively, first and second code generator circuits 18, 20 may include programmable floating gate transistors, or other programmed or programmable components which achieve the same operation.

Multiplexer circuitry 22 receives at data inputs thereof the outputs of signal generator circuits 18 and 20, i.e., the digital values corresponding to a first reference voltage Vref1 and a second reference voltage Vref2. Multiplexer circuitry 22 receives at a control and/or select input thereof the output of comparator 12. This output, representative of whether a relatively low temperature condition or relatively high temperature condition exists, is applied as a selection signal to multiplexer circuitry 22 so as to selectively place at the output of multiplexer circuitry. 22 a digital signal from circuit 18 corresponding to reference voltage Vref1 (if delta Vbe is to be measured and compared at the relatively low temperatures), or a digital signal from circuit 20 corresponding to reference voltage Vref2 (if Vbe is to be measured and compared at relatively high temperatures).

TTDC 10 further includes a digital-to-analog converter circuit 24 which receives the digital output of multiplexer circuit 22 and generates an analog voltage level Vref corresponding thereto. Converter circuit 24 may, for example, utilize a resistor voltage divider having switches, controlled by the digital output of the multiplexer circuit 22, which serve as taps along the resistor voltage divider. In this way, an analog voltage signal may be generated corresponding to the digital signal generated by the multiplexer circuit 22 and provided to converter circuit 24.

As shown in FIG. 2A, the output of converter circuit 24 is the analog reference voltage signal Vref which is applied to a second input of comparator 16. Again, this the analog reference voltage signal Vref may comprise reference voltage Vref1 (if delta Vbe is to be measured and compared at relatively low temperatures), or reference voltage Vref2 (if Vbe is to be measured and compared at relatively high temperatures). The comparator 16 functions to compare the analog reference voltage signal Vref to the output of multiplexer circuit 14 (delta Vbe or Vbe). When the voltage output of multiplexer circuit 14 (delta Vbe or Vbe) falls below the analog reference voltage signal Vref (Vref1 or Vref2), the comparator 16 output changes state thus indicating that the temperature environment of the IC or system 11 is extreme (too cold or too hot, respectively).

With reference to FIG. 2B, the output of converter circuit 24 is the analog reference voltage signal Vref which is scaled by circuit 15 (responsive to test mode signal TEST) and applied to a second input of comparator 16. Again, this the analog reference voltage signal Vref may comprise reference voltage Vref1 (if delta Vbe is to be measured and compared at relatively low temperatures), or reference voltage Vref2 (if Vbe is to be measured and compared at relatively high temperatures). The comparator 16 functions to compare the scaled analog reference voltage signal Vref to the output of multiplexer circuit 14 (delta Vbe or Vbe). When the voltage output of multiplexer circuit 14 (delta Vbe or Vbe) falls below the scaled analog reference voltage signal Vref (Vref1 or Vref2), the comparator 16 output changes state thus indicating that the temperature environment of the IC or system 11 is extreme (too cold or too hot, respectively). When not in test mode, the scaling circuit 15 is inactive and no change is made to the analog reference voltage signal Vref.

The operation of TTDC 10 will now be described. Initially, the output of comparator 12 identifies whether delta Vbe is to be compared to reference voltage Vref1 in order to determine if the temperature of the IC and/or system 11 falls below a predetermined low temperature, or whether Vbe is to be compared to reference voltage Vref2 to determine if the temperature of the IC and/or system 11 rises above a predetermined high temperature. This identification is reached by comparing delta Vbe to Vbe. If delta Vbe is greater than Vbe, a relatively higher temperature exists indicating that Vbe should be compared to reference voltage Vref2. See, FIG. 1A at reference 21. In this scenario, the output of comparator 12 is in a first logic state indicative of a relatively high temperature environment. Alternatively, if Vbe is greater than delta Vbe, a relatively lower temperature exists indicating that delta Vbe should be compared to reference voltage Vref1. See, FIG. 1A at reference 23. In this scenario, the output of comparator 12 is in a second logic state indicative of a relatively low temperature environment.

Next, multiplexer circuit 14 provides at its output a signal delta Vbe or Vbe, depending upon the logic state of the output of comparator 12. In other words, if a relatively low temperature condition exists (i.e., the output of comparator 12 is in the second logic state) and delta Vbe is to be compared with reference voltage Vref1, the delta Vbe signal is provided by multiplexer circuit 14 to comparator 16. If, on the other hand, a relatively high temperature condition exists (the output of comparator 12 is in the first logic state) and Vbe is to be compared with reference voltage Vref2, the Vbe signal is provided by multiplexer circuit 14 to comparator 16.

Further, multiplexer circuit 22 provides at its output either the digital signal corresponding to reference voltage Vref1 or the digital signal corresponding to reference voltage Vref2, based upon the logic state of the output of comparator 12. If the relatively low temperature condition exists and delta Vbe is to be compared with reference voltage Vref1, the digital signal corresponding to reference voltage Vref1 is provided by multiplexer circuit 22 to converter circuit 24. On the other hand, if the relatively high temperature condition exists and Vbe is to be compared with reference voltage Vref2, the digital signal corresponding to reference voltage Vref2 is provided by multiplexer circuit 22 to converter circuit 24. Converter circuit 24 converts the digital signal provided by multiplexer circuit 24 into an analog signal. The analog output of converter circuit 24 is analog reference signal Vref which is provided to comparator 16.

As a result, the analog signal Vref is a first analog voltage Vref1 to be compared with signal delta Vbe at relatively lower temperatures, and a second analog voltage Vref2 to be compared with signal Vbe at relatively higher temperatures. With signal generator circuits 18 and 20 providing output signals that are programmable, these first and second analog voltages may be defined according to customer requirements.

Comparator 16 compares analog signal Vref with the selected one of delta Vbe and Vbe signals. The alarm output of comparator 16, perhaps buffered by the flip flop FF, is used to signal whether an out-of-range (too high or too low) temperature exists and has been detected. Such a signal could be indicative of tampering with the IC or system 11. An IC or system 11 may, for example, use the output of comparator 16 to selectively take appropriate measures to protect or destroy the sensitive data stored in the system. The output of comparator 16 may be temporarily stored in the flip flop FF, the output of which is used to generate temperature detection alarm signal temp_alrm.

As explained above, TTDC 10 may be used to detect whether the IC is operating below a predetermined low temperature threshold and/or above a predetermined high temperature threshold (i.e., outside an expected operating temperature range). Voltage references Vref1 and Vref2, which correspond to the predetermined low and high temperature thresholds, respectively, may be programmed in order to allow for the predetermined low and high temperature thresholds to be defined according to system or customer requirements. This programming may occur by operating the IC in a test mode when subjected to the predetermined low and/or high temperature environment so as to determine the appropriate fuses to blow for generating voltage references Vref1 and Vref2.

The TTDC 10 may also include test circuitry for testing various aspects of TTDC 10 and for setting and/or programming the voltage references Vref1 and Vref2. Specifically, the test circuitry may include a counter 26 which generates a counter output digital signal coupled to a third data input of multiplexer circuit 22. When the IC or system 11 is placed in the test mode, and is at or beyond the designated threshold temperature, the multiplexer circuit 22 may be controlled by signal “test” to select and provide at its output the output of counter 22.

In order to program voltage reference Vref1 to correspond to the predetermined low temperature below which signal delta Vbe triggers the alarm signal generated by TTDC 10, the IC is exposed to the predetermined low temperature and placed in the test mode. Multiplexer circuit 22 is controlled using the “test” signal so as to pass to converter circuit 24 the output of counter 26. Multiplexer circuit 14 is controlled by comparator 12 so as to pass signal delta Vbe to comparator 16. Counter 26 is then activated to begin counting from an initial state. Converter circuit 24 converts the output of counter 26 into an analog signal which is then compared at each increment value to the delta Vbe signal by comparator 16. When the output of comparator 16 transitions from a first logic state to a second logic state, the incremented state of counter 16 is recorded (as being representative of the low threshold temperature) and used to determine a digital value and thus further identify the particular fuses in signal generator circuit 18 which need to be blown in order to generate a similar analog reference voltage to compare with delta Vbe. In the event a fuse implementation for circuit 18 is not used, the incremented state of counter 16 is recorded and used to determine a digital value that is to be programmed into the circuit 18.

Similarly, in order to program voltage reference Vref2 to correspond to the predetermined high temperature above which signal Vbe triggers the alarm signal generated by TTDC 10, the IC is tested at the predetermined high temperature and placed in the test mode. Multiplexer circuit 22 is controlled using the “test” signal so as to pass to converter circuit 24 the output of counter 26. Multiplexer circuit 14 is controlled so as to pass signal Vbe to comparator 16. Counter 26 is then activated to begin counting from an initial state. Converter circuit 24 converts the output of counter 26 into an analog signal which is then compared at each increment value to the Vbe signal by comparator 16. When the output of comparator 16 transitions from a first logic state to a second logic state, the incremented state of counter 16 is recorded (as being representative of the high threshold temperature) and used to determine a digital value and thus further identify the particular fuses in signal generator circuit 20 which need to be blown in order to generate a similar analog reference voltage to compare with Vbe. In the event a fuse implementation for circuit 20 is not used, the incremented state of counter 16 is recorded and used to determine a digital value that is to be programmed into the circuit 20.

In addition to using counter 26 to identify the fuses to blow (or programming to be made) in signal generator circuits 18 and 20, counter 26 may also be used to verify that the hysteresis of comparator 16 is operating correctly. In particular, once the output of comparator 16 transitions from the first logic state to the second logic state due to counter 26 counting in a first direction, such as incrementing, counter 26 may be further controlled to count in the reverse direction, such as decrementing. The hysteresis of comparator 16 may be successfully tested by counting in the reverse direction until the output of comparator 16 toggles back to the first logic state. By noting the state of counter 26 which caused the output of comparator 16 to toggle back to the first logic state, and comparing such counter state with the state of counter 26 which caused the output of comparator 16 to initially toggle from the first logic state to the second logic state, the existence of a hysteresis, and its magnitude, for the comparator 16 may be sufficiently tested.

A drawback of the foregoing procedure for test mode and programming of the voltages for vref1 and vref2 is that both of the too high and too low temperatures (i.e., temperatures outside of the expected operating temperature range) must be produced and the circuit 10 and/or IC or system 11 must be subjected to each of those temperatures during the test mode/programming phase. It may not be possible, or it may be inconvenient, to generate the too high and too low temperatures.

To address the foregoing drawback, the test circuitry within the TTDC 10 may further include circuitry for testing operation at a more convenient temperature (i.e., without subjecting the TTDC 10 or the IC or system 11 to extreme temperatures). For example, testing could be performed at or about room temperature (for example, at or about 20 degrees Celsius), which would be a temperature well within the expected operating temperature range. Any other suitable and conveniently obtainable temperature could alternatively be selected provided it also was within the expected operating temperature range.

Thus, according to an embodiment of the invention shown in FIG. 2A, TTDC 10 may include test circuitry 13 to test for correct high and low temperature operation using, for example, room temperature testing (or another suitably selected and conveniently obtainable temperature within the expected operating temperature range). This test circuitry 13 includes a scaling circuit 15 that receives the delta Vbe and Vbe signals. The scaling circuit 15 is operable to effectively change the received delta Vbe and Vbe signals during the test mode in a manner such that the delta Vbe and Vbe signals are each scaled to be at or substantially near the reference voltage Vref (i.e., within the Vref range as shown in FIG. 1A) when the TTDC 10 (or IC or system 11) is exposed to the suitably selected temperature (such as room temperature, for example, about 20 degrees Celsius). It will be noted that the reference voltage Vref is well below the supply voltage Vcc for the circuit 10.

With reference once again to FIG. 1A, the operation and effects of the scaling circuit 15 are shown. The received signals are the delta Vbe and Vbe signals. In one implementation, the scaling circuit 15 effectuates a voltage shift of the delta Vbe and Vbe signals to produce the delta Vbe′ and Vbe′ signals. It will be noted that, following the voltage shift (in the illustrated example a down-shift), both of the signals lie within the vref range (less than the supply voltage Vcc) at the suitably selected temperature within the expected operating temperature range (in this case, room temperature). Thus, this would allow, during test mode, for the determination of proper operation with respect to the setting and use of both of the programmed vref1 and vref2 values in comparison to the delta Vbe and Vbe signals without having to expose the circuit 10 or system or IC 11 to extreme temperatures.

FIG. 1A further shows another implementation operation of the scaling circuit 15. Again, the received signals are the delta Vbe and Vbe signals. In this implementation, the scaling circuit 15 effectuates a changing (i.e., flattening) of the slope of the substantially linear relationship between the delta Vbe and Vbe signals to temperature to produce the delta Vbe″ and Vbe″ signals. It will be noted that, following the flattening of the slopes, both of the signals lie within the vref range (less than the supply voltage Vcc) at the suitably selected temperature (in this case, room temperature). Thus, this would allow, during test mode, for the determination of proper operation with respect to the setting and use of both of the programmed vref1 and vref2 values in comparison to the delta Vbe and Vbe signals without having to expose the circuit 10 or system or IC 11 to extreme temperatures.

A multiplexer 17 receives the scaled delta Vbe′ and Vbe′ signals, or the scaled delta Vbe″ and Vbe″ signals, whichever are generated by circuit 15, and operates responsive to a test signal to select one of delta Vbe′/delta Vbe″ signals or one of the Vbe′/Vbe″ signals for output as the vtemp signal to comparator 16. That test signal may further control operation of the multiplexer 22 to select the counter 26 value for output to the converter 24. Alternatively, the test signal may control operation of the multiplexer 22 to select one or the other of the circuits 18, 20 to output the vref1 or vref2 signals to the converter 24.

In conclusion, following the scaling operation performed by the scaling circuit 15, signals delta Vbe and Vbe are scaled to be at or substantially near voltage reference Vref such that counter 26 and comparator 16 may be utilized as described above to see if the output of comparator 16 toggles at the expected state of the counter 26.

The operation for this test mode programming of the circuit 10 when exposed to a desired temperature within the expected operating temperature range would occur as follows.

The test circuitry of the TTDC 10 may include the counter 26 which generates a counter output digital signal coupled to a third data input of multiplexer circuit 22. When the IC or system 11 is placed in the test mode, and in this embodiment is placed at the desired temperature within the expected operating temperature range, the multiplexer circuit 22 may be controlled by signal “test” to select and provide at its output the output of counter 22.

The voltage reference Vref1 is now programmed to correspond to the predetermined low temperature below which signal delta Vbe triggers the alarm signal generated by TTDC 10. Multiplexer circuit 22 is controlled using the “test” signal so as to pass to converter circuit 24 the output of counter 26. Multiplexer circuit 17 is controlled by the test signals so as to pass a selected scaled signal delta Vbe to comparator 16 (for example, delta Vbe′ or delta Vbe″). Counter 26 is then activated to begin counting from an initial state. Converter circuit 24 converts the output of counter 26 into an analog signal which is then compared at each increment value to the delta Vbe′/delta Vbe″ signal by comparator 16. When the output of comparator 16 transitions from a first logic state to a second logic state, the incremented state of counter 16 is recorded (as being representative of the low threshold temperature) and used to determine a digital value and thus further identify the particular fuses in signal generator circuit 18 which need to be blown in order to generate a similar analog reference voltage to compare with delta Vbe. In the event a fuse implementation for circuit 18 is not used, the incremented state of counter 16 is recorded and used to determine a digital value that is to be programmed into the circuit 18.

Next, the voltage reference Vref2 is programmed to correspond to the predetermined high temperature above which signal Vbe triggers the alarm signal generated by TTDC 10. Multiplexer circuit 22 is controlled using the “test” signal so as to pass to converter circuit 24 the output of counter 26. Multiplexer circuit 17 is controlled so as to pass a selected scaled signal Vbe to comparator 16 (for example, Vbe′ or Vbe″). Counter 26 is then activated to begin counting from an initial state. Converter circuit 24 converts the output of counter 26 into an analog signal which is then compared at each increment value to the Vbe′/Vbe″ signal by comparator 16. When the output of comparator 16 transitions from a first logic state to a second logic state, the incremented state of counter 16 is recorded (as being representative of the high threshold temperature) and used to determine a digital value and thus further identify the particular fuses in signal generator circuit 20 which need to be blown in order to generate a similar analog reference voltage to compare with Vbe. In the event a fuse implementation for circuit 20 is not used, the incremented state of counter 16 is recorded and used to determine a digital value that is to be programmed into the circuit 20.

With respect to FIG. 2B, an alternative embodiment is presented for the test circuitry 13 to test for correct high and low temperature operation using, for example, room temperature testing (or another suitably selected and conveniently obtainable temperature within the expected operating temperature range). This test circuitry 13 includes a scaling circuit 15 that receives the analog reference voltage signal Vref which is output from the converter circuit 24. The scaling circuit 15 is operable, responsive to an asserted test mode signal TEST, to effectively change/shift the analog reference voltage signal Vref during the test mode in a manner such that the analog reference voltage signal Vref is scaled to be at or substantially near the delta Vbe and Vbe signals (as shown in FIG. 1B) when the TTDC 10 (or IC or system 11) is exposed to the suitably selected temperature (such as room temperature, for example, about 20 degrees Celsius). It will be noted that the reference voltage Vref (even when shifted) is set below the supply voltage Vcc for the circuit 10.

With reference once again to FIG. 1 B, the operation and effects of the scaling circuit 15 are shown. The received signal is the reference voltage Vref. The scaling circuit 15 effectuates a voltage shift of the reference voltage Vref. It will be noted that, following the voltage shift (in the illustrated example an up-shift), the delta Vbe and Vbe signals now lie with the range of the reference voltage Vref at the suitably selected temperature within the expected operating temperature range (in this case, room temperature). Thus, this would allow, during test mode, for the determination of proper operation with respect to the setting and use of both of the programmed vref1 and vref2 values in comparison to the delta Vbe and Vbe signals without having to expose the circuit 10 or system or IC 11 to extreme temperatures.

Operation of the FIG. 2B embodiment to set vref1 and vref2 values is performed in a manner analogous to that described above with respect to the FIG. 2A embodiment.

A possible implementation of TTDC 10 is shown in the schematic circuit drawings of FIGS. 3A-3L.

FIG. 3A shows a circuit diagram of a possible implementation of comparator 12. As discussed above, comparator 12 may have hysteresis (for example, an approximately 50 mv hysteresis). The comparator 12 has a structure which receives at differential inputs 51, 53 the delta Vbe and Vbe voltages. A differential comparison circuit 55 compares the signals at inputs 51, 53. The result of that comparison is a current which is mirrored in a current mirror circuit 57 to generate a logic signal at node 59 which is representative of the result of the comparison. The logic signal at node 59 is buffered through a pair of logic inverters to generate an output 61. A control circuit 63 responds to control inputs 64 to enable operation of the comparison circuit 55 and current mirror circuit 57.

FIG. 3B shows a circuit diagram of a possible implementation of converter circuit 24. However, while FIG. 2 shows signal generator circuits 18 and 20, counter 26 and multiplexer circuit 22 as being separate components from converter circuit 24, the implementation of the converter circuit 24 shown in FIG. 3B combines all those circuits together. As shown in FIG. 3B, the outputs y<7:0> of the counter 26 and the outputs tp<7:0> of the signal generator circuits 18, 20 (provided by, for example, a fuse register or other programmable circuitry) are provided to the first and second inputs, x<7:0> and y<7:0>, respectively, of the multiplexer circuit 22. The output z<7:0> of the multiplexer circuit 22 is provided to a first input x<7:0> of a hysteresis circuit 28. The hysteresis circuit 28 further receives at a second input hy<3:0> a digital signal indicative of a hysteresis value to be applied by the hysteresis circuit 28 to the signals provided at the first input x<7:0>. The digital signal applied to the second input hy<3:0> is obtained from a signal generator 34 (provided by, for example, a fuse register or other programmable circuitry) comprising hysteresis signal h<3:0>. The output z<7:0> of hysteresis circuit 28 is provided to the input trim<7:0> of a decoder 30, which provides a decoded digital signal b<15:0> to the input b<15:0> of a resistor array 32. The decoder 30 further generates from the input trim<7:0> the decoded control outputs msb, lsb, a<3:0> and abar <3:0> which are applied to corresponding inputs of the resistor array 32. The resistor array 32 responds to signals msb, lsb, b<15:0>, a<3:0> and abar <3:0> to perform a digital-to-analog conversion to generate of the signal trim<7:0> representing a reference voltage vref (such as the counter 26 output or the vref1 and vref2 outputs from the circuit 18, 20) to an analog value and thus generate an analog signal vout (i.e., analog vref) corresponding to the digital input received thereby. This analog output signal vout is then applied to one input of the comparator 16 (see, FIG. 2).

FIG. 3C shows a circuit diagram of a possible implementation of the counter 26. This counter 26 circuit is capable of both incrementing and decrementing, and thus can be used in test mode as described when incrementing to determine and set the digital values for the circuits 18, 20, and when decrementing for checking the hysteresis of the comparator 16. The counter 26 uses a plurality of logic gates (NANDs and NORs) along with 1 bit flip flops 27 or registers connected in a counter circuit configuration 29 for up- and down-counting with the count value output y<7:0> being taken at the output of each flip-flop 27. The input receives a control signal and the circuit counts up if the control signal is logic 1 and counts down if the control signal is logic 0.

FIG. 3D shows a circuit diagram of a possible implementation of signal generator circuits 18 and 20, each of which may include a number of fuse elements 81 which together form a fuse register. At each fuse location, a combination of an inverter 83 and transistor 85 forms a latch which operates to latch the lbit/hbit value which is indicated by the state of the fuse 81. The output of each fuse register lbit<7:0>/hbit<7:0> is a multi-bit digital signal that is provided to a multiplexer 38 which makes a selection between lbit<7:0> and hbit<7:0> to generate the output tp<7:0> which is provided to multiplexer circuit 22. Multiplexer 38 is controlled in part by the output of comparator 12. In circuit 18, the fuse register generates outputs lbit<7:0> for application to inputs x<7:0> of the multiplexer 38. In circuit 20, the fuse register generates outputs hbit<7:0> for application to inputs y<7:0> of the multiplexer 38.

FIG. 3E shows a circuit diagram of a possible implementation of multiplexer circuit 22. Responsive to the select signal, the circuit of FIG. 3E chooses either input x<7:0> or input y<7:0> to be transferred to output z<7:0>. To accomplish this goal, the x<7:0> and y<7:0> inputs are applied to complementary parallel connected pass transistors 87 whose gates are driven by complementary SELECT signals. Depending on SELECT signal state, one complementary parallel connected pass transistor 87 in a pair is actuated to pass its input to the output z<7:0>. A similar circuit may be used for the multiplexer 38 of FIG. 3D.

FIG. 3F shows a circuit diagram of a possible implementation of hysteresis circuit 28. The circuit receives signal x<7:0> and outputs signal y<7:0>. The circuit further receives a hysteresis offset value hy<3:0>. The hysteresis offset value hy<3:0> comprises hysteresis signal h<3:0> output from the hysteresis register 34 (see, FIG. 3B). Logic circuitry comprising a plurality of interconnected NAND gates 31 and half adder circuits 33 adds or subtracts the hysteresis offset value to the received signal x<7:0> to obtain the hysteresis adjusted output signal y<7:0>. A control signal add_sub controls operation of the logic circuitry to implement an add when the control signal is logic 0 and a subtract when the control signal is logic 1. It will be understood that the applied hysteresis could be zero in which case the hysteresis offset value hy<3:0> could be zero, or alternatively the hysteresis circuit 28 could be omitted from FIG. 3B completely by connecting the z<7:0> output of multiplexer 22 to the trin<7:0> input of decoder 30.

FIG. 3G shows a diagram for a possible implementation of decoder 30. The decoder includes a 4:16 decoder 43 (illustrated in the schematic of FIG. 3H) and a 2:4 decoder 45 (illustrated in the schematic of FIG. 31). The 4:16 decoder 43 receives the signals trim<4:1> from the hysteresis circuit 28 and decodes those signals to generate the signals b<15:0> for application to the resistor array. The signals b<15:0> control, in a manner to be described, resistor pass gates for taps in the resistor array for purposes of analog voltage selection. The 2:4 decoder receives the signals trim<6:5> and decodes those signals to produce signals b<19:16> for application to a logic circuit 41 which then generates the complementary control signals a<3:0> and abar<3:0> which control, in a manner to be described, resistor pass gates for shunts in the resistor array 32 with respect to course tuning of the generated analog voltage. The signals trim<0> and trim<7> are not decoded, but rather are buffered by two inverters and passed on to provide most significant bit (msb) and least significant bit (lsb) information which controls, in a manner to be described, resistor pass gates for shunts in the resistor array 32 for purposes of finest trimming of the generated analog voltage.

FIG. 3J shows a circuit diagram for a possible implementation of the resistor array 32 for use in performing the digital-to-analog decoding. As described above, this part of converter circuit 24 may include a resistor string (including a set of series connected resistors 322 forming a voltage divider) with a number of pass gate transistors 335 which effectively serve as taps in the string for providing a selected voltage from the resistors 322 of the resistor string for connection to the output vout. The pass gate transistors 335 are controlled by the digital signal b<15:0> generated by the decoder 30 from signal generator circuits 18, 20 and counter 26 to make a tap selection in the voltage divider and thus select a divider voltage for output. It will be noted that an inverter array 325 converts signals b<15:0> to bbar<15:0>, and that both b<15:0> and bbar<15:0> are used to control the pass gates of transistors 335 for the resistors 322 in tapping the voltage divider of the resistor string. In order to provide a suitable number of tap points along the voltage divider of the resistor string to provide selectable voltages across the entire supply voltage range without using an equivalent number of resistors and pass gate transistors, resistor array 322 may include additional resistors 321 which are connected in series with the resistor string. Resistors 321 may be selectively shorted (shunted) by pass gate transistors controlled by signals a<3:0> and abar<3:0> output from the decoder 30 from signal generator circuits 18, 20 and counter 26. Resistors 321 and the corresponding shorting pass gate transistors provide coarse tuning by effectively voltage shifting (up and down) the voltages along resistors 322 in the resistor string (voltage divider) that are coupled to output vout. As a result, fewer resistors 322 and corresponding pass gate transistors 335 need to be utilized to achieve the desired number of voltage levels for connection to output vout. The circuit further includes resistors 323 which are also connected in series with the resistors 322 of the resistor string. Resistors 323 may be selectively shorted (shunted) by pass gate transistors controlled by signals msb and lsb output from the decoder 30 from signal generator circuits 18, 20 and counter 26. Resistors 323 and the corresponding pass gate transistors provide finest trim tuning by effectively voltage shifting (up or down) the voltages along resistors 322 in the resistor string that are coupled to output vout.

The resistor string (including resistors 321, 322 and 323) is connected between a first voltage reference (Vcc1) and second voltage reference (ground).

FIG. 3J further illustrates that the resistor array has been duplicated for two different voltage levels (approx 1.2 v and approx 1.8 v). Thus a second resistor string (including resistors 321, 322 and 323) is connected between a third voltage reference (Vcc2) and fourth voltage reference (ground). In this implementation, the first voltage reference may be 1.2 v and the third voltage reference may be 1.8 v. The second and fourth voltage references may be ground.

A selection circuit 337 operates responsive to control signals to choose between the 1.2 v resistor string 322 and the 1.8 v resistor string 322 for connection to vout.

With reference to FIG. 3J-3, one possible implementation for the scaling circuit 15 (in this case with respect to the embodiment of FIG. 2B) is presented. The scaling circuit comprises a MOSFET device which is gate controlled by the test mode signal TEST. When TEST is asserted (active high), the MOSFET shorts the 0.6V point to ground. This increases the current in the resistor tree and provides a wider trimming range with a larger resolution. By doing this, when in test mode at room temperature, it is possible to cycle through the DAC covering a wider voltage window which would allow for a crossing of temp with Vref to occur at room temperature (while ordinarily, when not in test mode, this crossing would occur elsewhere in the temperature range).

FIG. 3K shows a circuit diagram for a possible implementation of a bandgap voltage generator. This circuit generates a reference current irefdvbe, a bias signal pbias_new and a bandgap output bgout. The circuit of FIG. 3K may further include a startup circuit in a manner known to those skilled in the art.

FIG. 3L shows a circuit diagram for a possible implementation of a signal generator circuit which includes one or more bipolar transistors possessing base-emitter junctions useful for measuring and monitoring Vbe and delta Vbe. The signal generator circuit receives the bandgap reference current Irefdvbe and bias signal pbias_new and operates to generate signals delta Vbe and Vbe. This circuit generates a 3 Vbe signal and dVbe signal. Three PNP transistor are stacked to form a 3 Vbe signal which has temperature coefficient of about −6 mv/c. Each PNP transistor is biased by pmos current source transistors with gate signal pbias_new. The DVBE3 signal is generated by feeding a PTAT (Proportional To Absolute Temperature, “Irefdvbe”) current into a resistor. The resulting DVBE3 voltage is thus also PTAT. The DVBE3 Temperature coefficient is about +6 mv/c. A PWRDWN signal is used to power down this circuit, by disconnecting the current source transistors. When PWRDWN goes HIGH, the transmission gate is shut off and disconnect the biasing current. It will be noted that Irefdvbe is generated in the bandgap (see, FIG. 3K), wherein a voltage across a resistor is developed which is the difference of two VBE voltages based on different current densities. The current density can be based on different sized emitters and different values of currents going through each bipolar transistor.

Thus, Vbe is measured by passing a current through the bipolar transistor. A voltage divider circuit includes a tap from which delta Vbe may be obtained. Output signals vbe2 and dvbe2 in FIG. 3L are input to comparator 12 in FIGS. 2 and 3A as the signals vbe and delta vbe, respectively. The output signals vbe3 and dvbe3 in FIG. 3L are indirectly coupled to comparator 16 of FIG. 2.

Although preferred embodiments of the method and apparatus of the present invention have been illustrated in the accompanying Drawings and described in the foregoing Detailed Description, it will be understood that the invention is not limited to the embodiments disclosed, but is capable of numerous rearrangements, modifications and substitutions without departing from the spirit of the invention as set forth and defined by the following claims. 

1. An integrated circuit temperature sensor, comprising: a scaling circuit operable to scale a voltage across the base-emitter of a bipolar transistor of the integrated circuit and a delta voltage across the base-emitter of a bipolar transistor of the integrated circuit to lie within a voltage reference range less than a supply voltage when the sensor is exposed to a temperature within an expected operating temperature range; a first multiplexer having first and second inputs coupled to receive the scaled voltage across the base-emitter of the bipolar transistor of the integrated circuit and the scaled delta voltage across the base-emitter of the bipolar transistor of the integrated circuit, the first multiplexer selecting one of the scaled voltages at the first and second inputs for output responsive to a mode signal; and a comparator operable to compare the voltage output from the first multiplexer against a reference voltage and generate an output signal responsive thereto.
 2. The sensor of claim 1 wherein the reference voltage comprises a selected one of a first reference voltage and a second reference voltage, and wherein the reference voltage is the first reference voltage if a first control signal is in a first state and wherein the reference voltage is the second reference voltage if a first control signal is in a second state.
 3. The sensor of claim 2 further including a second multiplexer having first and second inputs coupled to receive a first reference voltage signal and a second reference voltage signal, the first multiplexer selecting one of the signals at the first and second inputs for output as a reference voltage signal in response to the first control signal.
 4. The sensor of claim 3 wherein the reference voltage signal is a digital signal, the sensor further comprises a digital-to-analog converter operable to convert the digital reference voltage signal to an analog signal comprising the reference voltage.
 5. The sensor of claim 4 further comprising programmable circuitry defining the digital signal for the first reference voltage signal and the digital signal for the second reference voltage signal.
 6. The sensor of claim 5 wherein the programmable circuitry comprises fuse circuitry.
 7. The sensor of claim 4 further comprising: a counter operable to count a digital signal; and a circuit to selectively apply the digital signal to the digital-to-analog converter so as to provide a corresponding analog signal to the comparator.
 8. The sensor of claim 7 wherein the circuit to selectively apply comprises the second multiplexer operable responsive to the mode signal to select the digital signal instead of either the first reference voltage signal or the second reference voltage signal.
 9. An integrated circuit temperature sensing method, comprising: scaling a voltage across the base-emitter of a bipolar transistor of the integrated circuit and a delta voltage across the base-emitter of a bipolar transistor of the integrated circuit to lie within a voltage reference range less than a supply voltage while the sensor is exposed to a temperature within an expected operating temperature range; first selecting the scaled voltage across the base-emitter of the bipolar transistor of the integrated circuit if programming a high temperature threshold or selecting the scaled delta voltage across the base-emitter of the bipolar transistor of the integrated circuit if programming a low temperature threshold; and comparing the selected voltage against a reference voltage to generate an output signal.
 10. The method of claim 9 further comprising: varying the reference voltage; testing for a change in state of the output signal; and programming the high temperature threshold based on a value of the varying reference voltage at a point where the output signal changed state.
 11. The method of claim 9 further comprising: varying the reference voltage; testing for a change in state of the output signal; and programming the low temperature threshold based on a value of the varying reference voltage at a point where the output signal changed state.
 12. The method of claim 9 wherein the reference voltage is a digital signal, further comprising digital-to-analog converting of the digital reference voltage signal to an analog signal for use in the comparing.
 13. The method of claim 12 further comprising: varying the digital reference voltage signal; testing for a change in state of the output signal; and programming one of the high or low temperature thresholds based on a value of the varying digital reference voltage signal at a point where the output signal changed state.
 14. The method of claim 13 wherein programming comprises: applying a counter digital signal as the digital reference voltage signal, the counter digital signal incremented by the step of varying; and setting the value for the high or low temperature thresholds to a value of the counter digital signal at the point where testing indicates the change in state.
 15. An integrated circuit temperature sensor, comprising: a reference voltage generator operable in a first mode of operation to generate an incrementing reference voltage within a certain voltage range; a circuit operable during the first mode to scale a temperature dependent voltage signal concerning the base-emitter of a bipolar transistor of the integrated circuit and which is indicative of circuit operation outside of an acceptable temperature range to lie within the certain voltage range when the integrated circuit is exposed to a temperature that is within the acceptable temperature range; and a comparator operable to compare the scaled temperature dependent voltage signal against the incrementing reference voltage to generate an output signal and identify a certain incremented reference voltage at which the output signal changes state.
 16. The sensor of claim 15 further comprising: a reference voltage generator programmed for a second mode of operation to generate the certain incremented reference voltage, the comparator receiving the programmed certain incremented reference voltage during the second mode; and a circuit operable during the second mode to apply the non-scaled temperature dependent voltage signal concerning the base-emitter of a bipolar transistor of the integrated circuit to the comparator, a change in state of the output signal indicating that the circuit has been exposed to a temperature the outside of an acceptable temperature range.
 17. The sensor of claim 15 wherein the temperature dependent voltage signal concerning the base-emitter of a bipolar transistor of the integrated circuit comprises a voltage across the base-emitter of the bipolar transistor of the integrated circuit or a delta voltage across the base-emitter of the bipolar transistor of the integrated circuit.
 18. The sensor of claim 17 further a selection circuit operable to select in the second mode the delta voltage across the base-emitter of the bipolar transistor of the integrated circuit for comparison against the certain incremented reference voltage if the integrated circuit is currently exposed to the relatively low temperature or select in the second mode the voltage across the base-emitter of the bipolar transistor of the integrated circuit for comparison against the certain incremented reference voltage if the integrated circuit is currently exposed to the relatively high temperature.
 19. The sensor of claim 18 wherein the certain incremented reference voltage comprises a first reference voltage for use in comparison against the delta voltage across the base-emitter of the bipolar transistor of the integrated circuit or a second reference voltage for use in comparison against the voltage across the base-emitter of the bipolar transistor of the integrated circuit.
 20. An integrated circuit temperature sensor, comprising: a reference voltage generator operable in a first mode of operation to generate an incrementing reference voltage signal within a certain voltage range; a scaling circuit operable in the first mode to scale a temperature dependent voltage signal concerning the base-emitter of a bipolar transistor of the integrated circuit to lie within the certain voltage range less than a supply voltage when the sensor is exposed to a temperature within an expected operating temperature range; and a comparator operable to compare the incrementing reference voltage signal against the scaled temperature dependent voltage signal and generate an output signal responsive thereto.
 21. The sensor of claim 20 wherein the temperature dependent voltage signal is one of a voltage across the base-emitter of a bipolar transistor of the integrated circuit and a delta voltage across the base-emitter of a bipolar transistor of the integrated circuit.
 22. The sensor of claim 20 further comprising: a reference voltage generator programmed for a second mode of operation to generate a certain incremented reference voltage signal at which the output signal changed state in the second mode, the comparator receiving the programmed certain incremented reference voltage signal during the second mode; and a circuit operable during the second mode to apply the non-scaled temperature dependent voltage signal concerning the base-emitter of a bipolar transistor of the integrated circuit to the comparator, a change in state of the output signal indicating that the circuit has been exposed to a temperature the outside of the expected operating temperature range.
 23. An integrated circuit temperature sensor, comprising: a scaling circuit operable to selectively shift a voltage reference from a first range less than a supply voltage for one mode of operation to a second range less than a supply voltage during another mode of operation such that a voltage across the base-emitter of a bipolar transistor of the integrated circuit and a delta voltage across the base-emitter of a bipolar transistor of the integrated circuit lie within the second voltage reference range during the test mode of operation when the sensor is exposed to a temperature within an expected operating temperature range; a multiplexer having first and second inputs coupled to receive the voltage across the base-emitter of the bipolar transistor of the integrated circuit and the delta voltage across the base-emitter of the bipolar transistor of the integrated circuit, the multiplexer selecting one of the scaled voltages at the first and second inputs for output; and a comparator operable to compare the voltage output from the multiplexer against the shifted reference voltage and generate an output signal responsive thereto.
 24. An integrated circuit temperature sensing method, comprising: selectively shifting a voltage reference from a first range less than a supply voltage for one mode of operation to a second range less than a supply voltage during another mode of operation such that a voltage across the base-emitter of a bipolar transistor of the integrated circuit and a delta voltage across the base-emitter of a bipolar transistor of the integrated circuit to lie within the second voltage reference range during the another mode of operation when the sensor is exposed to a temperature within an expected operating temperature range; selecting one of the voltage across the base-emitter of the bipolar transistor of the integrated circuit and the delta voltage across the base-emitter of the bipolar transistor of the integrated circuit for output; and comparing the selected voltage against the shifted reference voltage to generate an output signal.
 25. The method of claim 24 further comprising: varying the reference voltage; testing for a change in state of the output signal; and programming a high temperature threshold based on a value of the varying reference voltage at a point where the output signal changed state.
 26. The method of claim 24 further comprising: varying the reference voltage; testing for a change in state of the output signal; and programming a low temperature threshold based on a value of the varying reference voltage at a point where the output signal changed state.
 27. An integrated circuit temperature sensor, comprising: a reference voltage generator operable to generate an incrementing reference voltage within a first voltage range less than a supply voltage; a circuit operable to selectively shift the incrementing voltage reference from the first voltage range to a second range less than a supply voltage such that a temperature dependent voltage signal concerning the base-emitter of a bipolar transistor of the integrated circuit and which is indicative of circuit operation outside of an acceptable temperature range lies within the second voltage range when the integrated circuit is exposed to a temperature that is within the acceptable temperature range; and a comparator operable to compare the temperature dependent voltage signal against the shifted incrementing reference voltage to generate an output signal and identify a certain incremented reference voltage at which the output signal changes state.
 28. The sensor of claim 27 wherein the temperature dependent voltage signal concerning the base-emitter of a bipolar transistor of the integrated circuit comprises a voltage across the base-emitter of the bipolar transistor of the integrated circuit or a delta voltage across the base-emitter of the bipolar transistor of the integrated circuit.
 29. The sensor of claim 1 wherein the mode signal is a test mode signal.
 30. The sensor of claim 15 wherein the first mode of operation is a test mode of operation for the integrated circuit.
 31. The sensor of claim 16 wherein the first mode of operation is a test mode of operation for the integrated circuit and the second mode of operation is a non-test mode of operation for the integrated circuit.
 32. The sensor of claim 20 wherein the first mode of operation is a test mode of operation for the integrated circuit.
 33. The sensor of claim 22 wherein the first mode of operation is a test mode of operation for the integrated circuit and the second mode of operation is a non-test mode of operation for the integrated circuit.
 34. The sensor of claim 23 wherein the one mode of operation is a non-test mode of operation for the integrated circuit and the another mode of operation is a test mode of operation for the integrated circuit.
 35. The sensor of claim 24 wherein the one mode of operation is a non-test mode of operation for the integrated circuit and the another mode of operation is a test mode of operation for the integrated circuit. 